Leakage Control

ABSTRACT

In one embodiment, a leakage reduction circuit is provided that includes: a virtual power supply node; a first PMOS transistor coupled between the virtual power supply node and a power supply node; a second PMOS transistor having a source coupled to the power supply node; and a native NMOS transistor coupled between a drain of the second PMOS transistor and the virtual power supply node, the native NMOS transistor having a gate driven by the power supply node.

RELATED APPLICATION

This application is a continuation-in-part of U.S. application Ser. No.12/144,622, filed Jun. 23, 2008, which is a continuation-in-part of U.S.application Ser. No. 11/857,133, filed Sep. 18, 2007, which is acontinuation of U.S. application Ser. No. 11/301,236, filed Dec. 12,2005, which claims the benefit of U.S. Provisional Application No.60/708,729 filed Aug. 16, 2005. In addition, this application claims thebenefit of U.S. Provisional Application No. 60/945,714, filed Jun. 22,2007 and also claims the benefit of U.S. Provisional Application No.60/990,967, filed Nov. 29, 2008.

The present invention relates to integrated circuits, and moreparticularly to integrated circuits having reduced leakage current.

As circuit dimensions continue to shrink, power dissipation due toleakage current is becoming an ever greater problem.Leakage-current-induced power dissipation in mobile devices such as cellphones reduces battery life, thereby inconveniencing users by requiringmore frequent re-charges. Ideally, a transistor in a digital integratedcircuit acts like a switch, being either in a conductive (on) state or anon-conductive (off) state. However, transistors always conduct someamount of leakage current in the off state. As process technologyadvances into the 90 nanometer (nm) or 65 nm dimensions and smaller, theability to close the channel between source and drain in a transistorweakens such that “subthreshold” leakage current continues to flowbetween the source and drain even when the transistor is turned solidlyoff.

Some approaches to mitigate subthreshold leakage current includelengthening the channel. However, that approach reduces achievablecomponent density, thereby obviating one of the major advantages ofmodern process technology. Rather than lengthen the channel, otherapproaches use multiple gates, which increases process complexity andstill reduces component density. Accordingly, there is a need in the artfor integrated circuits having improved leakage current reduction.

SUMMARY

This section summarizes some features of the invention. Other featuresare described in the subsequent sections.

In accordance with an aspect of the invention, a leakage reductioncircuit is provided that includes: a virtual power supply node; a firstPMOS transistor coupled between the virtual power supply node and apower supply node; a second PMOS transistor having a source coupled tothe power supply node; and a native NMOS transistor coupled between adrain of the second PMOS transistor and the virtual power supply node,the native NMOS transistor having a gate driven by the power supplynode.

The invention is not limited to the features and advantages describedabove. Other features are described below. The invention is defined bythe appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of an array of NMOS transistors with theirsources coupled to a virtual ground node controlled for leakage currentreduction.

FIG. 2 is a circuit diagram of an array of PMOS transistors with theirsources coupled to a virtual VDD node controlled for leakage currentreduction.

FIG. 3 is a circuit diagram of a virtual VDD node and a virtual groundnode configurably controlled for leakage current reduction.

FIG. 4 is a circuit diagram of an array of sub-arrays, wherein eachsub-array has a virtual ground controlled for leakage current reduction.

FIG. 5 is a circuit diagram of an array of X-decoders whose outputstages are selectively coupled to a virtual VDD node and a virtualground node for leakage current reduction.

FIG. 6 a illustrates a leakage reduction circuit in which transistor M2is implemented using a diode-connected native NMOS transistor that doesnot directly couple to ground.

FIG. 6 b illustrates a leakage reduction circuit in which transistor M2is implemented using a diode-connected native transistor that directlycouples to ground.

FIG. 6 c illustrates a leakage reduction circuit for a virtual powersupply node.

FIG. 7 illustrates an SRAM memory cell having a virtual ground node tocontrol leakage.

FIG. 8 illustrates a memory arranged into a plurality of sub-arrayshaving an access interface that controls the addressing of eachsub-array as well as leakage control modes.

DETAILED DESCRIPTION

Reference will now be made in detail to one or more embodiments of theinvention. While the invention will be described with respect to theseembodiments, it should be understood that the invention is not limitedto any particular embodiment. On the contrary, the invention includesalternatives, modifications, and equivalents as may come within thespirit and scope of the appended claims. Furthermore, in the followingdescription, numerous specific details are set forth to provide athorough understanding of the invention. The invention may be practicedwithout some or all of these specific details. In other instances,well-known structures and principles of operation have not beendescribed in detail to avoid obscuring the invention.

Conventional integrated circuits typically include circuit arrays inwhich only a few (typically just one) circuit is active at any giventime. For example, memories require address decoders such asrow-decoders or wordline-decoders wherein only a single decoder isactive depending upon the decoded address. Generally, such circuitsinclude a set of NMOS transistors sharing a common source node tied to alocal ground (VSS) and sharing a common drain node tied to VDD. The gatevoltages (Vg) of the NMOS transistors in the sets are driven to VSS ifthe corresponding circuit in which they are incorporated is inactive. Ifa circuit is active, however, Vg may either be held to VDD or switchedperiodically to VDD. Similarly, such circuits also include a set of PMOStransistors sharing a common source node tied to VDD and sharing acommon drain node tied to VSS. The gate voltages (Vg) of the PMOStransistors in a set in an inactive circuit are driven to VDD. If acircuit is active, however, Vg may either be held to VSS or switchedperiodically to VSS.

Given such a circuit array architecture, the potential for large amountsof leakage current is apparent. If the circuits are all inactive, theirsets of NMOS and PMOS transistors are turned off but havedrain-to-source voltages (Vds) equaling VDD. Thus, subthreshold leakagecurrent will flow through these sets of transistors. To stem this flowof leakage current, one or both sets of the NMOS and PMOS transistorsmay be modified to significantly reduce subthreshold and other types ofleakage current as will be explained further herein. For example,turning now to FIG. 1, consider the array of circuits 100 that includesets of NMOS transistors 105 operated as described above. For example,circuits 100 may be row-decoders, wordline-decoders, or any other typeof circuit arranged in an array such that one circuit (or a subset ofcircuits) is active at any given time. The sources of NMOS transistors105 are tied to a virtual ground 110 that couples to a local groundthrough NMOS transistors M1 and M2.

NMOS transistors M1 and M2 are controlled by mode signals V_(enable) andV_(standby), respectively. These mode signals may be assertedappropriately such that a normal mode, a standby mode, and a deep sleepmode of operation are enabled. The normal mode of operation correspondsto normal operation of circuits 100 (i.e, when one or more is active).In the normal mode, V_(enable) is asserted so that M1 is conducting. M1is sized so that it will easily supply the necessary current duringnormal operation. For example, M1 may be three times larger thantransistors 105. However, should all circuits 100 be inactive and M1still be conducting, the leakage current losses may be substantial. Toprevent such losses, Venable may be de-asserted when circuits 100 areinactive. However, to keep virtual ground 110 from floating too high(such that undesired delay would occur before M1 could drain the excesscharge on virtual ground 110 so that normal mode operation couldresume), V_(standby) may be asserted in a standby mode of operation ifall circuits 100 are inactive. M2 functions as a “chokepoint” to theleakage current flowing through transistors 105 during standbyoperation. In this fashion, M2 should be sized so as to be relativelyweak. For example, in one embodiment, if transistors 105 have a width tolength ratio (W/L) of 10/0.13 μm, M2 may have a W/L ratio of 0.3/2 μm.Leakage current from transistors 105 will cause a voltage rise atvirtual ground 110 during the standby mode because the leakage currentflow is being stymied through M2. It will be appreciated that if delayis not a concern, a more efficient design would exclude M2 and rely onM1 alone to reduce leakage current.

The voltage rise at virtual ground 110 will increase the current flowingthrough M2 in the standby mode. In a transistor such as M2 that isswitched on, the increase in current will be linearly proportional tothe increasing voltage at virtual ground 110. However, thegate-to-source voltage (Vgs) for transistors 105 becomes negative as thevoltage at virtual ground 110 increases because the gate voltage Vg fortransistors 105 is tied to the local ground, not virtual ground 110.Moreover, leakage is further reduced because the threshold voltage Vtfor transistors 105 is increased due to the body effect from the rise inpotential at virtual ground 110. It may be shown that the subthresholdleakage current through NMOS transistors decreases exponentially for anegative Vgs. Thus, the leakage current through transistors 105 willhave an exponential decrease in magnitude during the standby mode. Thesmaller the on-current for M2, the greater will be the rise in voltageat virtual ground 110 and thus the greater the increase in leakagecurrent reduction. However, the greater the rise in voltage at virtualground 110, the greater will be the time required to drain the excesscharge from virtual ground 110 if normal mode operation is desired.Circuits 100 typically cannot function properly if virtual ground 110 isnot virtually grounded during normal operation. Therefore, a tradeoffbetween leakage current reduction and time to recover for normal modeoperation governs the choice of a size (and hence on-current magnitude)for M2.

The deep sleep mode occurs if both V_(enable) and V_(standby) arede-asserted such that the virtual ground floats. In this fashion, theleakage current reduction is maximized. However, the delay necessary todrain excess charge from virtual ground 110 before normal operation maybe resumed from a deep sleep mode is greater as compared to a transitionfrom standby mode.

It may be observed that the leakage current reduction benefits providedby transistors M1 and M2 will be diminished if increasing numbers ofcircuits 100 are active in the normal mode of operation. For example,suppose all circuits 100 are active in the normal mode. Transistor M1must then be quite massive to supply the necessary current to theswitching transistors. To provide an equivalent leakage currentreduction using the die area space that would have to be devoted to M1,transistors 105 could simply have their W/L ratio adjusted to reducetheir leakage current. M2 need not be so massive, however, if only oneor a few circuits are active at any given time in the normal mode ofoperation.

The leakage current reduction explained with regard to NMOS transistors105 may be extended to sets of PMOS transistors 205 as shown in FIG. 2.An array of circuits 200 act as described with regard to circuits 100 ofFIG. 1 in that just a subset (such as just one) of circuits 200 isactive at any given time in a normal mode of operation. Each circuit 200includes one or more PMOS transistors 205 that have a common source nodeconnected to a virtual VDD node 210. If a circuit 200 is active, thegate voltage Vg for its corresponding PMOS transistors 205 is eitherheld low or is periodically switched low. However, if a circuit 200 isnot active, its gate voltage is pulled to VDD. Virtual VDD 210 couplesto actual VDD through PMOS transistors P1 and P2. P1 is the analog of M1(FIG. 1) in that it is sized sufficiently to conduct the necessarycurrent to PMOS transistors 205 in an active circuit 200. Because P1 isa PMOS transistor, its gate voltage V _(ENABLE) is the complement of thevoltage V_(enable) used to drive NMOS transistors 105 (should circuits200 include such transistors, for illustration clarity they are notillustrated). In a standby mode of operation in which all circuits 200are inactive, V _(ENABLE) is brought high. However, a gate voltage V_(STANDBY) for a P2 transistor is brought low in the inactive mode,where V _(STANDBY) is the complement of the voltage V_(standby) used todrive NMOS transistors 105 (if included in circuits 200). P2 is arelatively weak transistor sized analogously as discussed with regard toN2 to provide a chokepoint for the leakage current that would otherwiseflow through transistors 205. Because of the current flow through P2,virtual VDD will be slightly lower in potential than VDD in the inactivemode of operation. Because the gate voltage Vg applied to PMOStransistors 205 is true VDD (rather than virtual VDD), thegate-to-source voltage (Vgs) for PMOS transistors 205 is positive. Apositive Vgs voltage for PMOS transistors 205 has the same effect ofexponential leakage current reduction as does a negative Vgs voltage forNMOS transistors 105. In this fashion, the overall leakage currentthrough PMOS transistors 205 in the standby mode is substantiallyreduced. Further reduction of the leakage current is available if bothP1 and P2 are non-conducting in a deep sleep mode of operationanalogously as discussed with regard to FIG. 1.

The tradeoff governing the desired size for transistors N2 and P2 may beaffected by unpredictable or uncertain parameters. For example, leakagecurrent may be affected by semiconductor process variability (fast orslow process corners), temperature, and other variables that cannot bepredicted a priori. In turn, this unpredictability may complicate thedesign choice of how large transistors M2 and P2 should be to effect adesired tradeoff between leakage current reduction and recovery timenecessary to begin a normal mode of operation. To ease this designchoice, a series of selectable transistors N2′ through N2 ^(N) may beused in place of N2 and/or transistors P2′ through P2 ^(N) may be usedin place of P2 as seen in FIG. 3. The standby control signals would thenbe coded such that a suitable subset of these transistors conducts inthe standby mode. A controller (not illustrated) would determine theappropriate subset by selectively activating the transistors anddetermining whether the potential on virtual VDD 210 and virtual VSS 110in the standby mode is appropriate in regard to a desired tradeoffbetween leakage current reduction and recovery time.

Further leakage current reduction may be obtained by organizing thearray of circuits having its leakage current into groups of sub-arrays.For example, suppose circuits 100 comprise an array of 128 X-decoders.If a given X-decoder is active and the array is organized as discussedwith regard to FIGS. 1 and 2, all the remaining X-decoders may leakdespite not being used. However, if the X-decoders are arranged intosub-arrays, for example four sub-arrays of 32 X-decoders each, a givensub-array may be in the normal mode of operation while the remaininggroups are in the standby or deep sleep mode, thereby reducing leakagecurrent in the normal mode of operation. Moreover, each sub-array mayhave its own virtual ground and virtual VDD nodes as well. For example,turning now to FIG. 4, three sub-arrays 400-1 through 400-3 each have avirtual ground (elements 110-1 through 110-3, respectively), thatcouples to the sources of NMOS transistors (not illustrated) thatrequire leakage current reduction. Each virtual ground 110-1 through110-3 may be pulled low in a normal mode of operation by a correspondingNMOS transistor M1-1 through M3-1, respectively. Similarly, each virtualground 110-1 through 110-3 couples through an NMOS transistor M2-1through M2-3, respectively, to an NMOS transistor M3 controlled by thevoltage V_(standby). M3 thus functions as the analog of M2 in FIG. 1.Transistors M2-1 through M2-3 may either be tied to VDD as shown or tiedto the inverse of the corresponding V_(enable) signal.

In general, the larger the width a transistor 105 has as compared to agiven length, the greater its leakage current will be. Thus, there maybe a subset of NMOS transistors 105 that provide the great majority ofthe total leakage current. In that regard, the leakage reductioncircuits and techniques disclosed herein may be applied to just such asubset of transistors and achieve substantially the same overall leakagecurrent reduction. For example, turning now to FIG. 5, should circuits100 comprise X-decoders 500, it is conventional for each X-decoder 500to have relatively large (with regard to other transistors) transistorsin its output stage comprised of CMOS inverters 510. For example, afirst X-decoder 500-1 drives its output stages 510 to bring acorresponding word line X1 either high or low. Similarly, an nthX-decoder 500-n drives its output stages 510 to bring a word line XNeither high or low. The states of CMOS inverters 510 are known if allword lines are inactive. Each stage 510 will have either a PMOStransistor or an NMOS transistor that is turned off but conductingleakage current. For example, a first stage 510-1 has its input drivenhigh (corresponding to a logical ‘1’) by its corresponding X-decoder.Thus, a PMOS transistor 515 in each first stage 510-1 may be conductingleakage current if its X-decoder is inactive. To prevent this leakagecurrent, each PMOS transistor 515 has its source tied to a virtual VDDcontrolled, for example, as discussed with regard to FIG. 2. Similarly,each second stage 510-2 has its input driven low (corresponding to alogical ‘0’) by the corresponding first stage. Thus, an NMOS transistor520 may be conducting leakage current if its X-decoder is inactive. Toprevent this leakage current, each NMOS transistor 520 has its sourcetied to a virtual ground controlled, for example, as discussed withregard to FIG. 2. Finally, each third stage 510-3 has its PMOStransistor 530 controlled as discussed for stage 510-1.

The leakage current reduction techniques described herein may be appliedto two-dimensional arrays of memory circuits such as, for example, SRAMcells. In such arrays, it is conventional that only a single row isactive at any given time. Thus, each column in such an array has theproperty that only a single memory cell will be active at any giventime. Accordingly, each column in such an array may have its leakagecurrent reduced in accordance with the techniques disclosed herein. Forexample, in an SRAM cell, if the virtual VDD and virtual ground nodesare not allowed to float too far from VDD and VSS, respectively, thenthe memory contents of the SRAM cell will be preserved in the standbymode. However, the memory contents will eventually be lost in the deepsleep mode.

As discussed with regard to FIG. 3, process variations and other effectsmay make it difficult to predict in advance the particular sizetransistor best suited to serve as M2 or P2. For example, with regard toM2, it is desired to be weak such that it chokes the leakage current butyet not so weak that the virtual ground would raise too high inpotential. Regardless of its size, transistor M2 cannot conduct untilits threshold voltage Vt is exceeded. Once a non-native transistor suchas M2 is conducting, the relationship between the current it conductsand Vds is substantially linear. In general, the Vt for a non-nativetransistor is approximately 0.3 V, which means the virtual ground mustfloat up to this value before M2 conducts. However, should a processvariation be such that the total leakage current being conducted throughM2 is twice what is expected, the virtual ground could float toapproximately 0.6 V given the linear relationship between current andvoltage. Turning now to FIG. 6 a, it may be seen that forming transistorM2 using a diode-connected native transistors M2′ eases the designchoice. A native transistor has its channel blocked or masked during theimplant step that adjusts the threshold voltage for non-nativetransistors. In contrast to transistors with channel implants, an NMOSnative transistor has a threshold voltage equaling approximately 0volts. Moreover, the current is proportional to the square of the Vgsvoltage in a native NMOS transistor. Thus, even if process variationscause an unexpectedly high leakage currents to flow through M2′, thevoltage change at the virtual ground will be less dramatic as comparedto a non-native transistor embodiment. Because M2′ is diode connected,it must be switched through a serial connection to a transistor 600whose gate voltage is controlled by the V_(standby) as discussed withrespect to FIG. 2. Thus, when V_(standby) is asserted, the standby modeof operation is enabled. An alternative native transistor embodiment isshown in FIG. 6 b in which the native transistor is coupled betweentransistor 600 and ground. Such an arrangement is advantageous comparedto the embodiment of FIG. 6 a because the expected range of leakagecurrents is reduced.

An analogous diode-connected NMOS native transistor also simplifies thedesign of a virtual power supply VDD node as illustrated in FIG. 6 c. Arelatively strong PMOS transistor P1 couples between a power supply nodeVDD and the virtual power supply node VDD. An active low enable signalenx controls the gate of P1 such that when enx is asserted (broughtlow), the virtual power supply VDD node can supply power to circuits(not illustrated) that are powered through a coupling to this node. Incontrast, signal enx is de-asserted during a sleep mode such thatleakage control occurs through a diode-connected native NMOS transistorM1. Transistor M1 has its source connected to the virtual power supplyVDD node and its drain coupled to the drain of a PMOS transistor P2. Anactive low sleep signal controls the gate of transistor P2 such thatwhen the sleep signal is asserted, the native transistor M1 may conductthrough diode action since its gate is coupled to the power supply nodeVDD. In this fashion, the virtual power supply node VDD is kept slightlylower in potential than the power supply voltage VDD during the sleepmode while having the process variation resistance discussed with regardto FIGS. 6 a and 6 b. A deep sleep mode occurs when both enx and thesleep signals are high such that the virtual power supply node VDD mayfall so low in voltage that any active devices powered through this nodelose their necessary operating voltages.

The block-based leakage control discussed with regard to FIG. 4 leads toan advantageously simple control interface. For example, an array ofSRAM cells may be arranged in sub-arrays, wherein the leakage in eacharray is controlled on a block-by-block (sub-array) basis. As known inthe arts, each SRAM cell comprises a pair of cross-coupled invertersthat couple to a bit line and a complement bit line throughcorresponding access transistors. Turning now to FIG. 7, an SRAM cell700 thus includes cross-coupled inverters, where a first invertercomprises a PMOS transistor P1 that is coupled in series with an NMOStransistor M1. The second inverter comprises a PMOS transistor P2 thatis coupled in series with an NMOS transistor M2. An output node betweenP1 and M1 couples though an access transistor M3 to a bit line 710.Similarly, an output node 707 between P2 and M2 couples through anaccess transistor M4 to a complement bit line 710. Consider thesituation if node 701 is storing a logical one by being charged to apower supply voltage VDD such that complementary node 707 will begrounded. If a logical zero is to written into the cell, a word line(WL) coupled to the gate of access transistor M3 is activated while bitline 705 is grounded. M3 must then compete with P1 as to the voltagestate of node 701. Thus, it is conventional to make M3 larger than P1 sothat the memory cell may be “flipped” so as to change its binary state.Similarly, suppose the memory cell is storing a logical zero such thatnode 701 is grounded and a read operation is performed. The bit line 705will be pre-charged and then allowed to float while the word linevoltage is raised such that M3 conducts. Because of the pre-charge onthe bit line, there is a danger that node 701 will be momentarily raisedenough in voltage such that the memory cell changes its binary state. Toprevent such an error during read operations, it is conventional to makeM1 larger than M3. It may thus be seen that for a conventional SRAMcell, M1 and M2 represent the leakage problems since these transistorswill be considerably larger than P1 and P2. To prevent this leakage,transistors M1 and M2 couple to a virtual ground (denoted as VSS_cell)as discussed previously. A virtual power supply node such as discussedwith regard to FIG. 6 c could also be implemented but is typicallysuperfluous because of the relative sizes of P1 and P2 as compared to M1and M2. As discussed with regard to, for example, FIGS. 6 a and 6 b, arelatively large transistor M5 is controlled by a Venable signal that isasserted during normal operation (which may be denoted as an activemode). M5 is sized so as to provide the necessary current such that SRAMcell 700 retains its binary state during normal operation in which readand write operations are expected. A signal enx controls a gate of aPMOS transistor P4 coupled between a cell power supply node VDD_cell anda power supply node VDD. The cross-coupled inverter PMOS transistors P1and P2 have their sources tied to VDD_cell. Thus, during normaloperation, enx is asserted low to power cell 700.

In contrast to normal operation, a standby mode for cell 700 isestablished if no read or write operations are expected but cell 700should still retain its binary contents. To minimize leakage currentduring the standby mode, V_(enable) is de-asserted to turn off M5 whilea standby mode signal V_(standby) and enx are both asserted. The modesignal V_(standby) controls the gate of an NMOS transistor M6 that iscoupled in series to a diode-connected native transistor M7. In thisfashion, the diode-connected native transistor M7 controls the voltageof the VSS_cell node during the standby mode so as to prevent it fromfloating too high and destroying the binary contents of cell 700.However, the resulting negative Vgs on M1 or M2 (depending upon whichnode 701 or 707 is grounded) sharply reduces leakage current. It will beappreciated that M7 may be implemented as a non-native device dependingupon how much voltage raise is tolerated at VSS_cell.

A deep sleep (power down) mode is established for cell 700 if enx isde-asserted (brought to VDD) while V_(enable) and V_(standby) are bothde-asserted (brought low). Very little leakage occurs for such a modealthough the binary contents of cell 700 will be lost. The leakage willbe particularly minimized if P3 is constructed as a high Vt transistoror as an I/O transistor. Should an n-well for the memory array holdingcell 700 be connected to VDD_cell, there will be very little p+/nwelldiode leakage as well. A test mode may be enabled in addition to theactive, standby, and deep sleep modes. In the test mode, all leakagecontrol signals are turned off such that enx, V_(standby), V_(enable)are all asserted.

Turning now to FIG. 8, a memory 800 having a simple interface 801 isillustrated that uses the existing access interface for memorysub-arrays to control whether a given sub-array is active, in thelow-leakage mode (standby mode), or in a deep sleep mode. Memory 800includes three sub-arrays 805-1, 805-2, and 805-3. As known in the arts,an access interface 801 will decode an address so as to address a propersub-array. When a sub-array is accessed, it will trigger the assertionof a local clock in the sub-array that controls the decoding andlatching of a presented address. Suppose it is desired to place a givensub-array into the sleep mode. A sleep mode signal may be asserted thatis provided to a register 810 within each sub-array. A normal access tosub-array 805-1 is generated (either a read or a write operation). Inthis fashion, only the local clock within the addressed sub-array isasserted. However, in addition to the normal access to 805-1, a sleepenable signal trimload is asserted. Each sub-array includes an AND gate815 that provides the logical AND of its local clock and the trimloadsignal. Registers 810 are clocked by the output of the corresponding ANDgate such that only the register within the accessed sub-array isclocked. For example, suppose it is desired to place sub-array 805-1into the sleep mode. While an access operation (a dummy read or writeoperation to sub-array 805-1) occurs, sleep and trimload are asserted.Thus, only register 800 within sub-array 805-1 registers the sleepsignal, which is stored within this register as its Q signal. Subsequentdecoding circuitry (not illustrated) within each sub-array decodes the Qsignal such that the proper control (assertion of enx and de-assertionof V_(standby) and V_(enable)) occurs within the appropriate sub-array.A default mode for each sub-array would be the standby mode in whichV_(standby) is asserted while V_(enable) is de-asserted. During annormal read or write access to a sub-array, interface 801 assertsV_(enable) to allow normal operation in the accessed sub-array. To allowa universal sleep mode assertion for all sub-arrays, AND gate couldreceive an alternative signal to the local clock that represents alogical OR of the local clock and a universal sleep signal gpload. Thus,if trimload and gpload are both asserted, all the AND gates will triggertheir corresponding registers to latch the sleep signal. It will beappreciated the simplicity of modifying an existing address interface asdiscussed above to provide interface 801. For example, regardless of thenumber of sub-arrays within the memory, just two signals (sleep andtrimload) in conjunction with a normal access to a given sub-arraycontrol the assertion of the sleep mode.

The above-described embodiments of the present invention are merelymeant to be illustrative and not limiting. It will thus be obvious tothose skilled in the art that various changes and modifications may bemade without departing from this invention in its broader aspects.Therefore, the appended claims encompass all such changes andmodifications as fall within the true spirit and scope of thisinvention.

1. A leakage reduction circuit, comprising: a virtual power supply node; a first PMOS transistor coupled between the virtual power supply node and a power supply node; a second PMOS transistor having a source coupled to the power supply node; and a native NMOS transistor coupled between a drain of the second PMOS transistor and the virtual power supply node, the native NMOS transistor having a gate driven by the power supply node. 